Controller

ABSTRACT

A controller ( 1 ) comprises a comparator ( 10 ) which compares an input signal (Vo) with a reference signal (Vr) to obtain an error signal (ER). An integrator ( 11 ) applies an integrating action on the error signal (ER) to obtain a control signal (ICO). The integrator ( 11 ) allows influencing the integrating action. A copy circuit ( 81 ) supplies a copy control signal (ICOC) being proportional to the control signal (ICO). A determination circuit ( 85 ) determines whether the copy control signal (ICOC) reaches a limit value (IMIN, IMAX). An influencing circuit ( 83 ) influences the integrating action to limit the control signal (ICO) when the copy control signal (ICOC) reaches the limit value (IMIN, IMAX).

FIELD OF THE INVENTION

The invention relates to a controller, a current-mode controlled DC/DCconverter comprising such a controller, an apparatus comprising thecurrent-mode controlled DC/DC converter, and method of controlling.

BACKGROUND OF THE INVENTION

In a current-mode controlled DC/DC converter, a controllable switch iscoupled to an inductor to generate a periodically changing inductorcurrent through the inductor. An outer voltage regulation loop comprisesa current-mode controller that subtracts the output voltage of theconverter from a reference voltage to supply an error signal that isprocessed to obtain a control signal. This control signal may be used asa set level for the peak current in the inductor. The processing usuallycomprises a PI or a PID controller, which receives the error signal andsupplies the control signal. Therefore, often, this processing is alsoreferred to as a controller. An inner current regulation loop switchesoff the controllable switch when a sense signal that is representativefor the inductor current reaches the set level. Thus, the set level,which depends on the difference between the output voltage level and thereference voltage level, determines a peak current level of the currentthrough the inductor. Many options to determine this sense signal areknown. For example, the sense signal may be obtained with a currenttransformer, or as a voltage over an impedance in series with theinductor, this series impedance may be in the main current path of theswitch.

Usually, the switch is switched on by a clock pulse generated by anoscillator. The on-time of the switch is the period of time between theinstant the switch is switched on by the clock pulse and the instant theinductor current reaches the set level. The off-time of the switch isthe period in time between the instant the inductor current reaches theset level and the next clock pulse. The repetition period is the sum ofthe on-time and the off-time. In a buck converter, during the on-time,the switch connects the inductor between an input voltage and the outputand the inductor current increases. The input voltage may be supplied bya battery. During the off-time, another switch connects the inductorbetween the output and ground and the inductor current decreases. Thetopology of other current-mode controlled DC/DC converters, such as forexample, boost, buck-boost, Cuk converters, is also well known.

Usually, slope compensation is required to damp the disturbances in theinductor current. The slope compensation is obtained by varying the setlevel as a function of time during the repetition period. Often, thecurrent-mode controller either subtracts a sawtooth, a parabola, or apiecewise linear slope compensation signal from the control signal toobtain a slope compensated control signal. Now, this slope compensatedcontrol signal is used as the set level, and thus, the off-period startsat the instant the peak-current through the inductor reaches the levelof the slope compensated control signal.

In some applications, such as for example in telecom systems, thereference voltage is varied to obtain a varying output voltage whichfits the actual transmission power required. It is important that theoutput voltage of the power converter tracks the variations of thereference voltage optimally. It is a drawback of the known current-modecontrolled DC/DC converter that its speed of reacting on a variation ofthe reference voltage is not optimal.

U.S. Pat. No. 6,611,131 discloses such a current-mode switchingregulator. As prior art, a current-mode switching regulator is discussedin which a voltage clamp is present across the integrating capacitor ofthe I-controller. This voltage clamp limits the control voltage presentat the integrating capacitor. Thus, also the inductor current will belimited. It is further disclosed that this prior art solution has thedrawback that the actual value to which the inductor current is limiteddepends on the slope compensation. Therefore, U.S. Pat. No. 6,611,131proposes to use a voltage clamp which clamps the voltage at an output ofa buffer which buffers the voltage across the integrating capacitor. Thevoltage at which the output of the buffer is limited depends on theslope compensation. Now, the level at which the current is limiteddepends less on the slope compensation signal. However, this solutionhas the drawback that the integrating control loop is not closed duringthe limiting action and consequently, the voltage over the integratingcapacitor drifts away.

SUMMARY OF THE INVENTION

It is an object of the invention to provide a controller of which theoutput signal is limited and wherein the integrator drifts away less.

A first aspect of the invention provides a controller as claimed inclaim 1. A second aspect of the invention provides a current-modecontrolled DC/DC converter as claimed in claim 10. A third aspect of theinvention provides an apparatus as claimed in claim 16. A fourth aspectof the invention provides a method of controlling as claimed in claim18. Advantageous embodiments are defined in the dependent claims.

The controller in accordance with the first aspect of the inventioncomprises a comparator, an integrator, a copy circuit, a determinationcircuit, and an influencing circuit. Such a controller is usuallyreferred to as an I-controller. In the well known I-controllers, thecomparator compares an input signal with a reference signal and suppliesan error signal. The error signal indicates the difference between theinput signal and the reference signal. The integrator integrates theerror signal to obtain a control signal.

An example of the use of such an I-controller in a power converter isdisclosed in U.S. Pat. No. 6,611,131. In this application, the inputsignal of the I-controller is an output voltage of the power converter,and the reference signal is a reference voltage. The control signalsupplied by the I-controller controls the power converter such that itsoutput voltage is determined by the reference voltage.

The integrator allows influencing its integrating action. For example,if the integrator comprises an integrating capacitor, the integratorcomprises an input for influencing the current supplied to theintegrator capacitor. The copy circuit generates a copy control signalwhich is proportional to the control signal. The copy control signal maybe identical to the control signal but has to be present at a separatenode. The determination circuit determines whether the copy controlsignal reaches a limit value. The influencing circuit influences theintegrating action to limit the control signal when is determined thatthe copy control signal reaches or passes the limit value.

Thus, now, the separate circuit that compares the copy control signalwith the limit value is not disrupting the operation of the main pathwhich is formed by the comparator and the integrator. The loop from theinput signal to control signal is still fully operational, and thecontrol signal is still coupled to the state of the integrator. Or saiddifferently, by influencing the integrating action such that the controlsignal is limited, the link between the integrator state and the controlsignal is kept. In the prior art U.S. Pat. No. 6,611,131, the voltage atthe output of the buffer is limited, thus the limiting action takesplace in the main loop from input signal to control signal. However, thecontrol loop still detects a difference between the input signal and thereference signal and the integrator goes on acting to minimize thisdifference. Thus, the change of the integrator state is not anymorerepresented by the clamped output voltage of the buffer. Consequently,the integrator will drift far away from its nominal state. When thelimiting is not anymore required, it takes a long time before theintegrator changes back to its nominal state.

It has to be noted that the controller, besides the integrator,optionally, may comprise a proportional and/or a differentiating actionto obtain a PI, ID, or PID controller.

In an embodiment in accordance with the invention as claimed in claim 2the limit value indicates a maximum level, and the influencing circuitdecreases the integrating action when the copy control signal reachesthe maximum level. For example, if the integrating action is obtained bycharging a capacitor, this capacitor is discharged until the copycontrol signal is equal to the maximum level. And thus, because the copycontrol signal is a (scaled) copy of the control signal, also thecontrol signal is limited to a maximum value.

In an embodiment in accordance with the invention as claimed in claim 3the limit value indicates a minimum level, and the influencing circuitincreases the integrating action when the copy control signal reachesthe minimum level. For example, if the integrating action is obtained bycharging a capacitor, this capacitor is charged until the copy controlsignal is equal to the minimum level.

In an embodiment in accordance with the invention as claimed in claim 4,the copy circuit comprises a first current source which supplies thecopy control signal as a first current to a node. The determinationcircuit comprises a second current source that supplies a predeterminedfixed second current to the node which may be controllable. The firstcurrent and the second current have an opposite polarity to obtain adifference current. For example, the first current is drawn out of thenode while the second current is supplied to the node. The determinationcircuit further comprises a clamping circuit that limits a voltage atthe node. The influencing circuit comprises an amplifier that has aninput connected to the node and an output connected to the input of theintegrator to influence the integrating action. The difference currentflows into the clamping circuit as long as the voltage at the node is ina range wherein no clamping is required, the difference current flowstowards the influencing circuit if the voltage at the node reaches orcrosses the limit value at which limitation of the control signal isrequired. Such a controller with current sources can be easilyimplemented in an integrated circuit.

In an embodiment in accordance with the invention as claimed in claim 5,the integrator comprises an integrating capacitor, and the output of theinfluencing amplifier is coupled to the integrating capacitor to supplyor withdraw current from the capacitor when the control signal has to belimited. The amplifier has no influence at all if the control signalneed not be limited. Thus, if no limiting is required, the main loop isnot disturbed.

In an embodiment in accordance with the invention as claimed in claim 6,the integrator comprises a third current source to supply the controlsignal as a third current which is determined by a voltage across theintegrating capacitor. Now, also the control signal is generated with aneasy to integrate current source. It has to be noted that the voltage onthe integrating capacitor may control both the current sources forgenerating the control current and the copy control current.Alternatively, it is possible to only control the control current and tomirror the control current to obtain the copy control current.

In an embodiment in accordance with the invention as claimed in claim 7,the amplifier comprises a transistor, which has a control input coupledto the node, and a main current path coupled to the integratingcapacitor. Thus, the voltage on the node determines the amount ofcurrent supplied to or withdrawn from the integrating capacitor.

In an embodiment in accordance with the invention as claimed in claim 8,the clamping circuit comprises a transistor with a main current pathcoupled to the node and a control input. A voltage source is coupled tothe control input. Thus, depending on the arrangement of the transistorcontrol input and the level of the voltage source, the transistorconducts the difference current as long as the voltage at the node ishigher or lower than the voltage supplied by the voltage source.

In an embodiment in accordance with the invention as claimed in claim 9,the comparator comprises a first transconductance amplifier, which hasinputs to receive the input signal and the reference signal, and outputsto supply the error signal. Such a transconductance amplifier as such iswell known. The integrator comprises a capacitor arranged between theoutputs of the first transconductance amplifier. A secondtransconductance amplifier has inputs coupled across the capacitor, andoutputs to supply the control signal as an output current. The copycircuit comprises a third transconductance amplifier with inputs coupledacross the capacitor, and outputs to supply the copy control signal as acopy control current. The determination circuit comprises a currentsource arranged between the outputs of the third transconductanceamplifier, and a first transistor arranged between the outputs of thethird transconductance amplifier to act as a clamp. The current sourceis arranged to supply the limit value as a limit current. Theinfluencing circuit comprises a transistor with a main current patharranged between the outputs of the first transconductance amplifier,and a control input coupled to one of the outputs of the thirdtransconductance amplifier to influence a voltage across the capacitorwhen the copy control current reaches the limit current. The transistoris arranged to obtain a feedback loop. Such an implementation withtransconductance amplifiers is very suited to be used in an integratedcircuit.

In the current-mode controlled DC/DC converter in accordance with thesecond aspect of the invention, the controller is used to regulate thecurrent-mode controlled DC/DC converter (further also referred to as thepower converter). The power converter receives a power supply inputvoltage and supplies a power supply output voltage to a load. The powersupply output voltage is the input signal of the controller as mentionedearlier. The output signal of the controller, which is the controlsignal, is supplied to the power converter controller.

The power converter controller (also referred to as driver circuit)controls the power converter such that its output voltage is determinedby the control signal. In fact, in a current-mode controlled powerconverter, the control signal determines the level at which the currentin the inductor of the power converter reaches the limit value at whicha controllable switch coupled to the inductor is switched off. Thecontroller generates the control signal from the difference between theoutput voltage of the power converter and the reference voltage.

A driver circuit compares a sensed signal representative for theinductor current with the control signal to switch off the controllableswitch when a level of the sensed signal reaches a level of the controlsignal. Usually, the controllable switch is switched on by a clocksignal of an oscillator that is running on a fixed frequency. The on-and off-switching of the controllable switch causes a periodicallyvarying inductor current through the inductor.

The use of the controller in accordance with the invention allowslimiting the current through the inductor while the feedback loop fromthe output voltage of the power converter to the control signal is stillactive. This is because the link between the state of the integrator andthe control signal is still present. The extra path with the copycontrol current only influences the integrating action to obtain acontrol signal that is limited, but does not decouple the integratingaction and the control signal. For example, if the integrator is ananalog integrator that comprises an integrating capacitor, the voltageon the capacitor both represents the state of the integrator anddetermines the level of the control signal. Alternatively, if theintegrator is a digital integrator, the value reached by the integratorboth represents the state of the integrator and determines the level ofthe control signal. In fact, in accordance with the invention, the stateof the integrator is determined by the difference between the referencesignal and the input signal of the integrator. For example, if the inputsignal of the controller, which is the or a tapped in output voltage ofthe power converter, has a level higher than the level of the referencesignal, the integrator decreases the voltage on the capacitor or thevalue stored digitally, and the control signal decreases. The decreasingcontrol signal causes the driver to take actions to lower the outputvoltage of the power converter. In a current-mode power converterwherein the controllable switch is switched off when the sensed currentreaches the level of the control signal, the level of the controlsignal. should decrease if the output voltage of the power converter istoo high, such that the on-time of the switch decreases. In otherapplications this may be the other way around.

Usually, the I-controller subtracts the output voltage of the converterfrom the reference voltage to obtain the error signal. The controllerhas a transfer function that is applied on the error signal to obtainthe control signal. The transfer function, for example, may be anycombination of a P (proportional), I (Integrating), D (differentiating)regulator, but should have at least the I-action. Alternatively, thetransfer function may be a filter having at least an integrating action.Relevant is that the transfer function of the I-controller at leastcomprises an integrating action.

In an embodiment in accordance with the invention as claimed in claim11, the current-mode controlled DC/DC converter further comprises acorrection circuit which adds a correction signal to the control signalto obtain a modified control signal. The correction signal isrepresentative for a difference between the original control signal andan average value of the inductor current. A drive circuit compares asensed signal that is representative for the inductor current with themodified control signal to switch off the controllable switch when alevel of the sensed signal reaches a level of the modified controlsignal. Now, the switch is switched off when the level of the sensedsignal reaches the level of the modified control signal. Thus, thecontrol signal now more resembles the average value of the inductorcurrent. The limiting of the control signal now in fact advantageouslylimits the average inductor current instead of the peak inductorcurrent.

In the prior art current-mode controlled DC/DC converters, if no slopecompensation is present, the control signal is representative for thepeak level of the inductor current because the control signal determinesthe peak level of the inductor current at which the switch is switchedoff. If slope compensation is present, the slope compensated controlsignal still is representative for the peak level of the inductorcurrent. Consequently, the modified control signal is representative forthe peak level of the inductor current to which the slope compensationsignal is added. This is elucidated in detail with respect to FIGS. 6and 7. The open loop gain from the differential input voltage (thereference voltage level minus the output voltage level, or the other wayaround) to the output voltage depends on the topology of the controller,which now is also referred to as current-mode controller. Usually, thecurrent-mode controller is a P, a PI or a PID controller. The unity-gainfrequency of this open loop gain appears to depend on the transfer fromthe control signal to the average output current. In the prior art, thistransfer is smaller than 1 because the ripple current through theinductor causes the average inductor current to be smaller than the peakcurrent (the latter is controlled), and, if present, the slopecompensation also causes the peak inductor current to be smaller thanthe control signal.

In contrast, the current-mode controlled DC/DC converter in accordancewith the present embodiment of the invention comprises the correctioncircuit that receives the control signal and supplies a modified controlsignal which is used as the set level to be compared with the sensedlevel. The correction circuit adds a correction signal to the controlsignal to obtain the modified control signal. Because the modifiedcontrol signal still determines the peak level of the inductor current,now the control signal must be representative for the peak level of theinductor current minus the correction signal, if no slope compensationis present. Thus, if the correction signal is representative for thedifference between the peak inductor current and the average inductorcurrent, the control signal is more representative for the averageinductor current instead of the peak inductor current. Or said indifferent words, due to the closed loop from differential input voltageto the output voltage, at a same difference between the output voltageand the reference voltage, the modified control signal is independent onthe characteristics of the open loop from the differential input voltageto the set level. The output voltage has to reach the same value at asame peak value of the inductor current, and thus the set level (whichis now the modified control signal) should be the same. Consequently,the addition of the correction circuit which adds a correction signalrepresentative for the difference between the control signal and theaverage current through the inductor, causes the value of the controlsignal to drop with this difference. Now, the control signal supplied bythe current-mode controller is representative for the average inductorcurrent instead of the peak inductor current and/or slope compensationcurrent. The transfer function from the control signal to the averageoutput current becomes more equal to unity and the −3 dB bandwidthincreases. Further, advantageously, now the average inductor current islimited instead of the peak current.

The correction circuit may add a correction signal representative for adifference between an average value and an extreme value of the inductorcurrent. The control signal now becomes more equal to the averagecurrent through the inductor because the difference between the peakcurrent and the average current is compensated for. Or, at least, thisdifference is decreased.

Alternatively, if the current-mode controlled DC/DC converter furthercomprises a slope compensation circuit that generates a slopecompensation signal, again, the correction circuit adds the correctionsignal to the control signal to obtain a modified control signal. Now,the correction signal is, or is representative for, a sum of the levelof the slope compensation signal at the switch-off instant and thedifference between the peak current and the average current through theinductor. The difference between the peak current and the averagecurrent through the inductor was already catered for, the additionalattenuation introduced by the slope compensation is also removed.Consequently, the control signal is representative for the averagecurrent through the inductor. This again has the advantage that bylimiting the control signal of the controller, the average current ofthe power converter is limited, independent on the slope compensation.

In an embodiment in accordance with the invention as claimed in claim12, the limiting circuit is used to limit a minimum and/or maximum valueof the control signal. Now, because the control signal is representativefor the average current through the inductor, such a limiting circuitdirectly limits this average current. Not only a copy of the controlsignal but also a copy of the correction current is generated. Thelimiting circuit now detects when the sum of the copy control signal andthe copy correction signal reaches the limit value, and influences theintegrator accordingly to obtain the limited control signal.

In an embodiment in accordance with the invention as claimed in claim13, again, easy to integrate current sources are used to implement thecopy control signal and the clamping circuit.

In an embodiment in accordance with the invention as claimed in claim14, further, the copy correction signal is generated by an easy tointegrate current source.

In an embodiment in accordance with the invention as claimed in claim15, the current-mode controlled DC/DC converter is a buck converter, andthe signals are currents that are summed at a node. The current-modecontroller comprises a controlled current source, which supplies acontrol current determined by the control signal to the node. Thecorrection circuit comprises a current source, which supplies thecorrection signal as a correction current to the node. A sense circuitsenses the inductor current and supplies the sensed signal as a sensedcurrent to the node. The polarities of the control current and thecorrection current are the same are and opposite to a polarity of thesensed current. Thus, if, for example, the sensed current flows towardsthe node, both the control current and the correction current flow awayfrom the node. The driver circuit is coupled to the node to determinewhen the level of the sensed current crosses the level of the sum of thecontrol current and the correction current. If the sensed currentcrosses this sum, the switch is switched off.

These and other aspects of the invention are apparent from and will beelucidated with reference to the embodiments described hereinafter.

BRIEF DESCRIPTION OF THE DRAWINGS

In the drawings:

FIG. 1 shows a block diagram of a prior art current-mode controlledDC/DC converter,

FIG. 2 shows a circuit diagram of an embodiment of the controller inaccordance with the invention,

FIG. 3 shows a circuit diagram of an embodiment of the controller inwhich the control signal is limited to a maximum value,

FIGS. 4 show signals elucidating the limitation of the control signal toa maximum value,

FIG. 5 shows a circuit diagram of an embodiment of the controller inwhich the control signal is limited to a minimum value,

FIG. 6 shows a circuit diagram of an embodiment of the current-modecontrolled DC/DC converter in accordance with an aspect of theinvention,

FIG. 7 shows signals elucidating the operation of the prior artcurrent-mode controlled DC/DC converter,

FIG. 8 shows signals elucidating the operation of the current-modecontrolled DC/DC converter shown in FIG. 6,

FIG. 9 shows a circuit diagram of an embodiment of the controller of thecurrent-mode controlled DC/DC converter in which the modified controlsignal is limited to a minimum value,

FIG. 10 shows a circuit diagram of an embodiment of the controller andthe correction circuit for implementation in an integrated circuit.

FIG. 11 shows a block diagram of embodiment of the current-modecontrolled DC/DC converter in accordance with the invention, and

FIG. 12 shows a block diagram of another embodiment of the current-modecontrolled DC/DC converter in accordance with the invention.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT

FIG. 1 shows a block diagram of a prior art current-mode controlledDC/DC converter. Especially in telecom systems wherein a handheld has tomanage the transmission-power economically to increase the battery life,the power supply voltage of a transmitting output amplifier should becontrolled to optimally suite the actual transmission power. Thecurrent-mode DC/DC converter (further also referred to as powerconverter, or converter) which supplies the power supply voltage shouldbe able to modulate its output voltage fast and accurate. In the nowfollowing, the current-mode DC/DC converter is also referred to as powerconverter or even as converter. In this application in a handheld, theload LO represents circuits of the handheld, such as the outputamplifier.

The power converter comprises current-mode controller 1 (also referredto as the controller) which supplies a control signal ICO which dependson a difference between the output voltage Vo of the converter and areference voltage Vr. The reference voltage Vr may be constant or may bevaried to obtain a corresponding varying output voltage Vo. The currentmode-controller 1 comprises a subtractor 10 which subtracts the outputvoltage Vo from the reference voltage Vr to supply an error signal ERwhich represents the difference between the reference voltage Vr and theoutput voltage Vo. Instead of the subtractor 10 also a comparator, whichcompares the output voltage Vo with the reference voltage, Vr to obtainthe error signal ER may be used. The current-mode controller 1 furthercomprises an integrator 11 which integrates the error signal ER toobtain the control signal ICO. Usually, the controller 1 is a P(Proportional) controller, an I (Integrating) controller, a PI(Proportional and Integrating) controller, or a PID (Proportional,Integrating and differentiating) controller. Such controllers are wellknown in the art. If in the now following is referred to the controller1, it is meant that the controller at least comprises an integratingaction performed by the integrator 11. However, the controller 1 mayfurther also comprise a proportional and/or differential action.

The optional slope compensation circuit 2 subtracts a slope compensationsignal from the control signal ICO to obtain a slope compensated controlsignal SCO. Usually, the slope compensation signal is sawtooth shaped,parabola shaped, or piecewise-linearly shaped. A sense circuit 6 sensesthe current IS1, which flows through the switch S1. The sense circuit 6may sense any current which is representative for the inductor currentIL through the inductor L. For example, the sense circuit 6 may bearranged in series with the inductor L to sense the inductor current ILdirectly, or the sense circuit 6 may be arranged in series with theswitch S1 (as shown) or in series with the switch S2. If the sensecircuit 6 is arranged in series with one of the switches S1 or S2, theinductor current IL is only sensed during the period in time theassociated switch is closed. The sense signal SE which should berepresentative for the inductor current IL may also be sensed as avoltage, for example this voltage may be sensed over the main currentpath of one of the switches S1 or S2. Preferably, the switches S1 and S2are MOSFET's, but bipolar transistors or other controllablesemiconductor devices may also be used. Instead of the switch S2, adiode may be used.

The comparator 3 compares the sensed signal SE with the slopecompensated control signal SCO to supply a reset signal RS to the resetinput R of the Set-Reset Flip-Flop 4 when the level of the sensed signalSE reaches the level of the slope compensated control signal SCO.Instead of the Set-Reset Flip-Flop 4 a more complicated circuit may beused. An oscillator 5 generates a clock signal CLK which is supplied tothe set-input S of the Set-Reset Flip-Flop 4. The non-inverting output Qof the Set-Reset Flip-Flop 4 supplies a control signal SC1 to a controlinput of a switch S1, and the inverting output Qn of the Set-ResetFlip-Flop 4 supplies a control signal SC2 to a control input of a switchS2. However, usually, the control of the synchronous switch S2 may bemore complicated. It is also possible that the switch S2 is a diode.Then, of course, no control signal is required. When the Set-ResetFlip-Flop 4 is reset by the reset signal RS of the comparator 3, theswitch S1 is opened and the switch S2 is closed. When the Set-ResetFlip-Flop 4 is set by a clock pulse CLK on the set input S, the switchS1 is closed and the switch S2 is opened.

The main current paths of the switch S1 and S2 are arranged in seriesbetween the terminals of a DC power supply which supplies the inputvoltage Vb to the converter. An inductor L is arranged between thejunction of the main current paths of the switch S1 and S2 and theoutput of the converter where the output voltage Vo is supplied. Aparallel arrangement of a smoothing capacitor C and a load LO is presentat the output of the converter. The current through the inductor isindicated by IL.

The operation of the prior art buck converter is briefly elucidated inthe now following. It is assumed that the starting situation is that theclock pulse CLK sets the Set-Reset Flip-Flop 4. Now, the switch S1 isclosed and the switch S2 is opened causing the inductor current IL toincrease. The inductor current IL increases until the sensed signal SEis equal to the compensated control signal SCO. Now the Set-ResetFlip-Flop 4 is reset by the reset signal RS, the switch S1 is opened andthe switch S2 is closed. The inductor current IL decreases untilSet-Reset Flip-Flop 4 is set again by a next clock pulse CLK.

FIG. 2 shows a circuit diagram of an embodiment of the controller inaccordance with the invention. The current-mode controller 1 elucidatedwith respect to FIG. 1, now is called the controller 1 which performs atleast an integrating action. The controller 1 comprises the comparatoror subtractor 10, the integrator 11, the copy circuit 81, thedetermination circuit 85, and the influencing circuit 83. The comparatoror subtractor 10 determines an error signal ER which is the differenceof the reference signal Vr and the input voltage Vo which should becontrolled. The integrator 11 integrates the error signal ER andsupplies the control signal ICO. The integrator 11 has an input IINwhich allows influencing the integrating action of the integrator 11.The controller 1 has an input at which the input signal Vo is received,and an output at which the control signal ICO is supplied. The inputsignal Vo is compared with the reference signal Vr and the difference ERis integrated to, supply the control signal ICO which can be used tocontrol a further circuit such that the input signal Vo becomes equal tothe reference signal Vr.

The copy circuit 81 receives the control signal ICO and supplies a copycontrol signal ICOC that is proportional to the control signal ICO. Forexample, if the control signal ICO is a current, this current may bemirrored to obtain the copy. The determination circuit 85 receives thecopy control signal ICOC, and supplies a control signal LCS indicatingwhether the copy control signal ICOC reaches or trespasses a limit valueIMIN or IMAX. The influencing circuit 83 receives the control signal LCSto supply an influencing signal IA to the input IIN of the integrator 11to influence the integrating action of the integrator 11. Withinfluencing the integrating action is, for example, meant that an extrasignal is added to the error signal ER such that the integrator suppliesa corrected control signal ICO. For example, if the determinationcircuit 85 detects that the copy control signal ICOC reaches the maximumlevel IMAX, the influencing circuit 83 generates an influencing signalwhich decreases the error signal ER, or which is subtracted from theerror signal ER such that the control signal ICO does not furtherincrease. Effectively, when limiting, due to the closed loop formed bythe copy circuit 81, the determination circuit 85, and the influencingcircuit 83, the integrating action will be influenced by the influencingsignal IA such that the copy control signal ICOC is limited to themaximum level IMAX. A similar reasoning holds for limiting the copycontrol signal ICOC to a minimum level.

Because the limiting action is applied on the copy control signal ICOCand not on the control signal ICO, the loop from the input voltage Vo tothe control signal ICO is not opened. Consequently, the control signalICO and the state of the integrator 11 are still in conformance witheach other. Thus, when the limiting action is no longer required, thestate of the integrator 11 has not drifted away, and the time requiredfor the integrator 11 to retrieve its nominal state is not excessivelylong.

It has to be noted that by limiting the copy control signal ICOC, alsothe control signal ICO is limited. In regulation loops in which thecontroller 1 is used, often there is a need to limit the control signalICO, for example to prevent an overload in the circuit to be controlled.Especially when the circuit to be controlled is a power converter, thecurrents through or the voltages over components of the power convertershould be limited. In other applications, the speed with which thecircuit is controlled may have to be limited.

It has to be noted that the main loop, which comprises the subtractor 10and the integrator 11, is a known I-controller. The copy circuit 81, thedetermination circuit 85, and the influencing circuit 83, which arecollectively also referred to as the limiting circuit 8, define anembodiment of the present invention.

FIG. 3 shows a circuit diagram of an embodiment of the controller inwhich the control signal is limited to a maximum value. The controller 1comprises the integrator 11 which comprises an integrating capacitor C1and the current source 112. In FIG. 3 the controller 1 only comprisesthe integrator 11 and thus is an I-controller. The voltage VC across thecapacitor C1 controls the current source 112 to supply the controlsignal ICO which is a current. However, the controller 1 may alsocomprise a P-action and/or a D-action (both not shown), which alsoinfluence the current source 112, and thus the control signal ICO. Theintegrating action of the integrator 11 may also by obtained with adigital solution.

The voltage VC across the capacitor C1 is supplied both to the currentsource 112 to obtain the control current ICO and to the current source81 of the limiting circuit 8 to obtain a copy control current ICOC ofthe control current ICO. Alternatively, the voltage VC may only besupplied to the current source 112, and the copy control current ICOCmay be a mirrored and/or scaled version of the control current ICO. Thecopy current ICOC is drawn out of a node N2. The limiting circuit 8further comprises a current source 80, a clamping circuit 82 and anamplifier 83. The current source 80 supplies the current IMAX to thenode N2. The current IMAX represents the maximum value to which the copycurrent ICOC should be limited. The clamping circuit 82 is coupled tothe node N2 to limit the voltage VN at the node N2 to a maximum value.The difference current ICL is equal to the current IMAX minus the copycontrol current ICOC. The input of the amplifier 83 receives the voltageVN and its output is connected to the input of the integrator 11 todecrease the integrating action of the integrator 11 if the copy currentICOC reaches or surpasses the current IMAX. In the embodiment shown,wherein the integrator 11 comprises an integrating capacitor C1, theamplifier 83 draws a current IA out of this capacitor C1 when thelimiting is active.

FIG. 3, as an example only, shows a particular embodiment of theclamping circuit 82 and the amplifier 83. The clamping circuit 82 andthe amplifier 83 are designed such that always only one of them conductscurrent. The clamping circuit 82 comprises a FET 820 that has a maincurrent path arranged between the node N2 and a reference potential,which in FIG. 3 is ground. A voltage source 821 that supplies a voltagelevel VCLH is connected between the control electrode of the FET 820 andthe reference potential. The amplifier 83 comprises a FET 830 that has acontrol electrode connected to the node N2, and a main current pathconnected between the input Il of the integrator 11 and the referencepotential. As long as the copy current ICOC is smaller than the maximumcurrent IMAX, the clamping circuit 82 sinks the difference current ICLand limits the voltage VN to a maximum value and the amplifier 83current IA is zero. As soon as the copy current ICOC becomes larger thanthe maximum current IMAX, the difference current TCL changes polarityand the voltage VN drops. Due to the decreased level of the voltage VN,the clamping circuit 82 stops sinking current, and the amplifier 83starts to draw current from the capacitor C1 to decrease the integratingaction. The control loop, created by the limiting circuit 8 when theamplifier 83 is active, is designed to have a large open loopamplification factor such that the integrating action is influenced toobtain a copy current ICOC which is limited to the maximum current IMAX.Consequently, also the control current ICO is limited to a maximumvalue. The operation of the limiting circuit 8 of FIG. 3 is elucidatedin more detail with respect to FIGS. 4.

FIGS. 4 show signals elucidating the limitation of the control signal toa maximum value. FIG. 4A shows the differential input voltage Vr−Vo, orthe error signal ER which is the input signal of the integrator 11. FIG.4B shows the voltage VC on the capacitor C1 of the integrator 11. FIG.4C shows the copy control current ICOC and the control current ICO. Itis assumed that the copy current ICOC is equal to the control currentICO. However, in a practical implementation, the copy control currentICOC may be a scaled version of the control current ICO. FIG. 4D showsthe difference current ICL, FIG. 4E shows the voltage VN at the node N2,and FIG. 4F shows the current IA drawn out of the integrating capacitorC1.

It is assumed that the controller is operating in an open-loop mode andthat at the instant t0 the level of the differential input signal Vr−Vois increased. The integrator 11 starts charging the capacitor C1 and thevoltage VC starts increasing. Although in FIG. 3 the integrator 11 onlycomprises an integrating action, it is in the now following assumed thatthe integrator 11 is a PI-controller. The control current ICO and itscopy ICOC show a proportional increment (indicated by P in FIG. 4C), andan integrating increment (indicated by I in FIG. 4C). The differencecurrent ICL is flowing towards the clamping circuit 82, the voltage VNis high and thus the clamping circuit 82 is able to sink the decreasingdifference current ICL. The difference current ICL decreases because theincreasing copy control current ICOC comes nearer to the maximum currentIMAX supplied to the node N2 by the current source 80. The current IAconducted by the amplifier is zero due to the high level of. the voltageVN.

At the instant t1, the copy control current ICOC becomes equal to themaximum current IMAX. Now, the difference current ICL becomes zero orslightly negative and the voltage VN drops to a low level. Consequently,the clamping circuit 82 stops conducting and the amplifier 83 startsconducting the current IA. Now a feedback loop is obtained. Theamplifier 83 has a large current gain, and the input current of theamplifier 83′ is negligible, thus equilibrium in the feedback loop isrestored when the copy control current ICOC becomes equal to the maximumcurrent IMAX. Thus, the copy control current ICOC is limited to themaximum value IMAX.

At the instant t2, the differential input voltage Vr−Vo is furtherincreased. The proportional part of the controller 11 outputs a higherproportional current in the control current ICO and its copy ICOC. Thisadditional current is not shown in FIG. 4C because it will beimmediately compensated by the compensating action of the amplifier 83which increases the current IA to compensate for the extra proportionalcurrent. This extra current IA is obtained by a further decrease of thevoltage VN at the node N2.

At the instant t3, the reference voltage is decreased such that theinput difference voltage Vr−Vo becomes negative. The proportional partof the integrator 11 outputs a negative proportional contribution P′ inthe control current ICO and its copy ICOC which values now immediatelydrop below the maximum value IMAX. The voltage VN quickly rises, theamplifier current IA stops flowing, and the clamping circuit 82 startsconducting the increasing difference current ICL. The current limitingloop is now opened and the voltage VC on the capacitor C1 is not anymoreinfluenced by the limiting circuit 8. Due to the negative inputdifference voltage Vr−Vo, the voltage VC on the capacitor C1 startsdecreasing. The integrating part of the integrator 11 is indicated byI′.

It has to be noted that, although the limiting circuit 8 is elucidatedwith respect to an analog integrator 11 with the capacitor C1, theintegrator 11 may also be implemented with digital circuits such as acounter. The amplifier 83 now has to act on the up-down countingmechanism of the counter. The integrator 11 may also lack the P-actionand/or may include a D-action.

If the controller 1 is implemented in a power converter, it is importantto select the value of the maximum current IMAX such that the limitingcircuit 8 limits the control current ICO before the protection of themaximum current through the transistor switch S1 is activated, andbefore the inductor L is saturated.

It has further to be noted that existing protection circuits which haveto protect the switches S1 and S2 of the power converter against toolarge currents, are unable to limit the average output current IOA ofthe converter. Instead, they limit the maximum current through theswitches because of the presence of the ripple current. However, theripple current varies with the output voltage. The ripple currentamplitude is maximal when the output voltage is approximately half thebattery voltage Vb, and the ripple current amplitude is approaching zerofor output voltages near zero volt or near the battery voltage Vb.

A first known protection circuit, senses the current through the controlswitch S1 and compares it which a maximum value. The control switch S1is immediately reset when is detected that the current through thecontrol switch S1 becomes larger than the maximum value. The controllerwill respond with increasing the control current, and the next switchingcycle, again the control switch S1 will be immediately reset when isdetected that the current through the control switch S1 becomes largerthan the maximum value. This will last until the cause of the too largecurrent is taken away. In fact, the limiting loop opens the main loop,and thus it will take a considerable amount of time to recover from anover-current state.

As discussed earlier, U.S. Pat. No. 6,611,131 discloses two other priorart protection circuits which have the associated problems mentionedhereinbefore.

FIG. 5 shows a circuit diagram of an embodiment of the controller inwhich the control signal is limited to a minimum value. The controller 1is identical to that shown in FIG. 3. Thus, as in FIG. 3, the voltage VCacross the capacitor C1 is supplied to the current source 112 to obtainthe control current ICO, and to the control source 81 of the limitingcircuit 8 to obtain a copy control current ICOC of the control currentICO. Again, the copy control current ICOC is drawn out of the node N2.

The limiting circuit 8 further comprises a current source 80′, aclamping circuit 82 and an amplifier 83. The current source 80′ suppliesthe current IMIN to the node N2. The current IMIN represents the minimumvalue to which the copy current ICOC should be limited. The clampingcircuit 82 is coupled to the node N2 to limit the voltage VN at thisnode to a minimum value. The input of the amplifier 83 receives thevoltage VN and its output is connected to the input of the integrator 11to supply a current IA to the capacitor C1 to increase the integratingaction if the copy current ICOC reaches or drops below the current IMIN.

The operation of the limiting circuit 8 shown in FIG. 5 is comparable tothat of the limiting circuit 8 shown in FIG. 3. Briefly, as long as thecopy control current ICOC is larger that the minimum current IMIN, thevoltage VN at the node N2 is low and the difference current ICL isconducted by clamping circuit 82. The amplifier 83 is inactive and -thecurrent IA is zero. When the copy current ICOC becomes equal to theminimum current IMIN, the voltage VN increases and causes the clampingcircuit 82 to stop conducting current. The amplifier 83 starts supplyingthe current IA into the capacitor C1 to prevent the copy current ICOC tofurther decrease.

FIG. 5, as an example only, shows a particular embodiment of theclamping circuit 82 and the amplifier 83. The clamping circuit 82 andthe amplifier 83 are designed such that always only one of them conductscurrent. The clamping circuit 82 comprises a FET 822 which has a maincurrent path arranged between the node N2 and a reference potential,which in FIG. 3 is a positive potential V+. A voltage source 823, whichsupplies a voltage level VCLL, is connected between the controlelectrode of the FET 822 and another reference potential (ground). Theamplifier 83 comprises a FET 831 which has a control electrode connectedto the node N2, and a main current path connected between the input IIof the integrator 11 and the reference potential V+. As long as the copycurrent ICOC is larger than the minimum current IMIN, the clampingcircuit 82 sources the difference current ICL and limits the voltage VNto a minimum value. As soon as the copy current ICOC becomes equal tothe minimum current IMIN, the difference current ICL changes polarityand the voltage VN rises. Due to the increased level of the voltage VN,the clamping circuit 82 stops sourcing current, and the amplifier 83starts to supply current to the capacitor C1 to increase the integratingaction. The control loop, created by the limiting circuit 8 when theamplifier 83 is active, is designed to have a large open loopamplification factor such that the integrating action is influenced toobtain a copy current ICOC which is limited to the minimum current IMIN.Consequently, also the control current ICO is limited to a minimumvalue.

FIG. 6 shows a circuit diagram of an embodiment of the current-modecontrolled DC/DC converter in accordance with the invention. Thisembodiment is based on the block diagram of the prior art convertershown in FIG. 1. FIG. 6 shows a possible implementation in an integratedcircuit that uses current sources.

First, the circuit equivalent with the converter shown in FIG. 1 isdiscussed, the current source 70 which supplies the correction currentICR is assumed to be not yet present. The controller 1 comprises thesame subtractor 10, which receives the reference voltage Vr, and theinput voltage Vo (which is the output voltage of the power converter) tosupply the same error signal ER. The integrator 11 integrates the errorsignal ER to obtain a control signal CO which controls the currentsource 111 to draw the control current ICO from the node N1. Thecurrent-mode controller CC may, besides the integrator 11 also comprisea P and/or D action to obtain a PI, or PID controller to generate thecontrol signal CO from the error signal ER.

The slope compensation circuit 2 comprises a current source 20 whichsupplies a slope compensation current IS1 to the node N1. The sensecircuit 6 now supplies a sensed current ISE, which is representative forthe inductor current IL, to the node N1. A voltage at the node N1 isdetermined by the sum of the currents ICO, ISE and ISL. The comparator 3now comprises the amplifier 30 which supplies the reset signal RS whichindicates when the level of the sensed current ISE becomes equal to thedifference of the control current ICO and the slope compensation currentISL. Both the oscillator 5 and the Set-Reset Flip-Flop 4 are identicalto the same items shown in FIG. 1. Also the topology formed by theswitch S1, S2, the inductor L, the capacitor C and the load LO isidentical to that shown in FIG. 1. The operation of this implementationin an integrated circuit of the known converter, and the drawbacksthereof are elucidated in detail with respect to the signals shown inFIG. 7.

In an embodiment of the converter in accordance with the presentinvention, a correction circuit 7 is added. In the embodiment shown inFIG. 6, the correction circuit 7 comprises a current source 70 whichdraws a correction current ICR out of the node N1. The operation of thisembodiment is elucidated in detail with respect to the signals shown inFIG. 8. Alternative embodiments of the correction circuit 7 arediscussed with respect to FIGS. 11 and 12.

FIG. 7 shows signals elucidating the operation of the prior artcurrent-mode controlled DC/DC converter. FIG. 7 shows a steady statesituation wherein the level of the inductor current IL at the end t=T ofa switching period T is identical to the level of the inductor currentIL at a start t=0 of a switching period T. The current IS1 through theswitch S1 is identical to the inductor current IL during the on-periodlasting from the instant 0 to the instant DT during which the switch S1is closed. The sensed current ISE is proportional to the current IS1through the switch S1. The control current ICO has a predeterminedconstant level in the steady state. The difference current of thecontrol current ICO and the slope compensation current ISL is shown asthe curve indicated by ICO-ISL. At the instant DT, the sensed currentISE becomes equal to the difference current ICO-ISL and the Set-ResetFlip-Flop 4 is reset. The switch S1 is opened and the switch S2 isclosed. Now, during the off-period which lasts from the instant DT tothe instant T, the inductor current IL decreases. The current IS1through the switch S1 and thus the sensed current ISE drop to zero, theslope compensation current ISL is switched off (ISL=0) and thedifference current ICO-ISL becomes equal to the control current ICO. Ithas to be noted that in a practical embodiment the currents may bescaled versions of the real currents. The average inductor current ILAis indicated by the dashed line. In a buck converter, the average outputcurrent IOA is the current supplied to the parallel arrangement of thesmoothing capacitor C and the load LO and thus is equal to the averageinductor current ILA. This average output current IOA is averaged overthe switching period T. For a boost converter, which has a switch S2 atits output, the current supplied to this parallel arrangement differsfrom the average current ILA through the inductor L.

From FIG. 7 it becomes clear that the gain from the control current ICOto the average output current IOA is not 1. This is caused by the slopecompensation current ISL and the ripple IRI on the inductor current IL.The slope compensation current ISL causes the control current ICO to belarger than the peak inductor current ILP. The ripple current IRI causesthe average inductor current ILA to be lower than the peak inductorcurrent ILP. The gain Ai from the control current ICO to the averageoutput current IOA isAi=IOA/ICOTo elucidate the effect on the small-signal bandwidth of thecurrent-mode controlled DC-DC buck converter, it is assumed that thecontroller 11 is a PI-controller, such that the transfer from the inputs(Vr and Vo) to the output (ICO) of the current-mode controller 1 isICO/(Vr−Vo)=gHF*(1+jωτ)/jωτwherein gHF is the value of the high-frequency transfer (theproportional part), and τ is the time constant of the integrating part.The output voltage Vo is filtered by the capacitor C, and the load LO isconsidered to be a resistor. Therefore, the transfer from the averageoutput current IOA to the output voltage Vo isVo/IOA=R/(1+jωRC)The open loop gain from the differential input voltage Vr−Vo to theoutput voltage Vo is thusVo/(Vr−Vo)=Ai*gHF*R*(1+jωτ)/(jωτ*(1+jωRC))The open loop gain has a low-frequency pole at fp=1/(2jπRC) and ahigh-frequency zero at fz=1/(2πτ).The unity-gain frequency of the open-loop gain isfl=(Ai*gHF)/(2πC)The closed-loop gain has a −3dB bandwidth f3 that can be approximated bythe unity-gain frequency fl of the open-loop. Thus, the closed-loop −3dBbandwidth f3 depends on the value of the output capacitor C, thehigh-frequency transfer gHF, and the gain Ai of the transfer fromcontrol current ICO to average output current IOA. The values of thecapacitor C and the transfer gHF are well known, however, the value ofthe gain Ai is smaller than 1, and is not fixed. Due to the fact that Aiis smaller than 1, the closed-loop bandwidth of the transfer fromreference voltage Vr to output voltage Vo is smaller than maximalpossible. This is a disadvantage because it limits the possibilities ofthe converter to accurately follow fast variations of the referencevoltage Vr at the output.

FIG. 8 shows signals elucidating the operation of the current-modecontrolled DC/DC converter shown in FIG. 6. Now, the correction circuit7 has been added which comprises a current source 70 which draws acorrection current ICR out of the node N1. Because, in the same steadystate, the same current IS1 flows through the switch S1, the sensedcurrent ISE is identical to that of the prior art converter. Also, theslope compensation current ISL is considered to be identical to that ofthe prior art converter. Again, in the same steady state, the totalcurrent at the node N1 should cause a reset of the Set-Reset Flip-Flop 4at the same instant DT. Consequently, the effect of adding thecorrection circuit 7 is that the control current ICO must decreaseexactly with the value of the correction current ICR.

Thus, if the correction current ICR is selected to be equal to the sumof the level of the slope compensation current ISL at the switch offinstant DT and half the ripple current IRI, the control current ICObecomes equal to the average inductor current ILA. Consequently, thegain Ai of the transfer from control current ICO to average outputcurrent IOA becomes equal to 1 and the closed-loop bandwidth of thetransfer from reference voltage Vr to output voltage Vo has its maximumvalue.

In the now following, the operation of the converter, which has such acorrection circuit 7, is elucidated. Again, by way of example only, theconverter is a buck converter, and the controller CC is a PI controllerwhich comprises the integrator 11. Further, the current source 70 drawsthe correction current ICR from the node N1, by way of example, near tothe current source 111 which draws the control current ICO from the nodeN1, as shown in the Figure. The sum of the correction current ICR andthe control current ICO is denoted by the sum-current IMC which is drawnout of the node N1. The sum of the slope compensation current ISL andthe sensed current ISE is flowing towards the node N1. Thus theSet-Reset Flip-Flop 4 will be reset at the instant DT that the sensedcurrent ISE reaches the level of sum-current IMC from which the slopecompensation current ISL is subtracted. The sum-current IMC is alsoreferred to as the modified control signal MCO in FIGS. 11 and. 12.

In FIG. 8, it is assumed that the correction current ICR has a valuesuch that the modified control signal IMC has the same level as thecontrol signal ICO in FIG. 7. Consequently, the control signal ICO inFIG. 8 corresponds directly with the average inductor current ILA andthe average output current IOA. The word “corresponds” is used toindicate that scaled versions of the actual currents may be used. In allother aspects, FIG. 8 is identical to FIG. 7.

In the now following calculation, the value of the correction currentICR is determined for a buck-converter wherein the slope compensation isparabola shaped. From FIG. 7 it follows that the difference between thecontrol current ICO and the average output current IOA isICO−IOA=ISL(DT)+IRI/2wherein ISL(DT) is the slope compensation current at the instant DT atwhich the switch S1 is switched off, and IRI is the peak to peak ripplecurrent through the inductor current IL. The optimal slope compensationcurrent ISL for a buck-converter isISL(t)=½*(t/T)²*(T/L)*Vb=(t ² Vb)/2TLwherein t/T is the relative position in a clock cycle with duration T, Lis the inductor value of the inductor L, and Vb is the DC input voltageof the converter. This input voltage may be supplied by a battery.At the instant of switching off the switch S1 (which is also referred toas the control switch), the slope compensation current ISL has the valueISL(DT)=½*D ²*(T/L)*Vbwherein D is the steady state value of the duty cycle, which, ifneglecting losses, is Vo/Vb. The peak to peak ripple current on the coilcurrent ILA or the output current IOA is for a buck-converterIRI=DT*(Vb−Vo)/LWith the above equations, the difference between the control current ICOand the average output current IOA isICO−IOA=ISL(DT)+IRI/2=(T*Vo)/(2L)Consequently, if the correction current ICR has this value (T*Vo)/(2L),the control current ICO becomes equal to the average inductor currentILA and thus also to the average output current IOA. It has to be notedthat the correction current ICR is a positive feedback current.

The current gain Ai which describes the transfer from the controlcurrent ICO to the average output current IOA, now has a unitymagnitude. Consequently, the −3dB bandwidth f3 of the loop has increasedtof3≈gHF/(2πC).A further advantage is that the 31 3dB bandwidth depends on twowell-known quantities only.

A similar improvement of the reaction speed is obtained if otherconverter topologies than a buck-converter are used, or when the PIcontroller CC has another behavior, or when the slope compensation has adifferent shape or is not present at all.

FIG. 9 shows a circuit diagram of an embodiment of the current-modecontrolled DC/DC converter in which the modified control signal islimited to a minimum value larger than zero. FIG. 9 is based on FIG. 5,the first difference is that a current source 70 which conducts thecorrection current ICR has been added at the output of the currentsource 112, as also is shown in FIG. 6. The sum of the correctioncurrent ICR and the control current ICO is the modified control currentIMC. The second difference is that a current source 71 has been added atthe node N2 to conduct a copy current ICRC of the correction currentICR. The sum of the copy correction current ICRC and the copy controlcurrent ICOC is the modified copy current IMCC.

The clamping circuit 82 conducts current as long as the modified copycurrent IMCC as larger than the minimum current IMIN which is largerthan zero. The amplifying circuit 83 is inactive and thus does notinfluence the integration node in the integrator 11. When the modifiedcopy current IMCC becomes smaller than the minimum current IMIN suppliedby the current source 80′, the clamping circuit 82 ceases conduction andthe amplifier 83 starts supplying current IA to the capacitor C1 of theintegrator 11. Consequently, the copy control current ICOC is controlledsuch that the modified copy current IMCC is limited to the level of theminimum current IMIN.

To determine the proper value of the minimum current IMIN, first theoperation is considered of the circuit of FIG. 6, but without thecorrection circuit 7, as is elucidated with respect to FIG. 7. In thisprior art circuit, the reset input R of the Set-Reset Flip-Flop 4becomes active (high) at the instant DT at which the current through thecontrol switch S1 becomes equal to or larger than a difference controlcurrent ICO-ISL which is equal to the control current ICO minus theslope compensation current ISL, see FIG. 7. As a result, the controlswitch S1 becomes non-conductive and the slope compensation currentsource 20 is switched off. In order to be sure that the reset input R ismade inactive (low), it is required that the difference control currentICO-ISL is at least larger than the sensed current ISE which ispositive.

Now, it is assumed that further the correction current source 70, whichsupplies the correction, current ICR is present in the FIG. 6 topology.The difference between the control current ICO and the modified controlcurrent IMC is equal to the correction current ICR. Again, to be surethat the reset input R is made inactive the modified control current IMCshould be positive. Consequently, the minimum current IMIN should beselected larger than zero.

A consequence of meeting the requirement that the modified copy currentIMCC cannot become smaller that the minimum current IMIN is that theaverage inductor current ILA may become negative. The converter is ableto convert the energy stored in the smoothing capacitor C back to thepower supply voltage Vb. The converter now more or less operates as aboost converter from the output capacitor C to the battery whichsupplies the supply voltage Vb. It has to be noted that the current inthe switch S1 now may become negative, thus this switch S1 should havebi-directional current capability. Also the switch S2 should havebi-directional current capability and thus should be a synchronousswitch and not a diode.

FIG. 10 shows a circuit diagram of an embodiment of the controller andthe correction circuit implemented in an integrated circuit. Anattractive manner to realize a PI-controller in an integrated circuit isto use fully differential circuitry to profit maximally from common-moderejection to suppress spurious signals, which are often present,especially in switched-mode power supplies. The required common-modecontrol loops, which set the common mode voltages of the nodes toappropriate values, are not shown.

The transconductance amplifier TCA3 receives the reference voltage Vr atthe non-inverting input and the output voltage Vo at the invertinginput, and supplies output currents to nodes N3 and N4. Thetransconductance amplifier TCA3 has a transfer determined by thetransconductance gHF which represents the high-frequency proportionalpart of the PI-controller 1 or CC. The low-frequency integrating part ofthe PI-controller is generated by the transconductance amplifiers TCA1and TCA2 and the capacitor C1. These amplifiers TCA1, TCA2, and thecapacitor C1 thus form the integrator 11. The transconductance amplifierTCA1 with transconductance gLF1 receives the reference voltage Vr at thenon-inverting input and the output voltage Vo at the inverting input,and supplies output currents to the capacitor C1. The transconductanceamplifier TCA2 with transconductance gLF2 receives the voltage acrossthe capacitor C1 between the non-inverting input and the invertinginput, and supplies its output currents to the nodes N3 and N4. As faras the components discussed so far are considered, an attractiveIC-implementation of the prior art PI-controller is obtained. The sum ofthe currents at the nodes N3 and N4 form the output currents indicatedby IMC. These currents IMC form the control signal ICO of FIG. 1.

The control signal ICO may be changed into the modified control currentIMC which corresponds to the modified control current IMC shown in FIG.6 by adding the transconductance amplifier TCA4 with transconductancegCOR. The transconductance amplifier TCA4 has a non-inverting inputwhich receives the output voltage Vo and an inverting input which isconnected to a reference voltage which is ground. The transconductanceamplifier TCA4 supplies the correction currents ICR to the nodes N3 andN4.

The limiting circuit 8 which limits the maximum value of the controlsignal ICO comprises: the current source 80 which supplies the maximumcurrent IMAX from the node N5 to the node N6, the transconductanceamplifier TCA5 with transconductance gHF, the transconductance amplifierTCA6 with transconductance gLF2, and the FET's F1 and F2. In fact, F2 ispart of the amplifier 83. The transconductance amplifier TCA5 receivesthe reference voltage Vr at the non-inverting input and the outputvoltage Vo at the inverting input, and supplies output currents to thenodes N5 and N6. The transconductance amplifier TCA6 receives thevoltage across the capacitor C1 between the non-inverting input and theinverting input, and supplies its output currents to the nodes N5 and N6also. Thus, the transconductance amplifier TCA5 supplies theproportional part of the copy control current ICOC of FIG. 3, and thetransconductance amplifier TCA6 supplies the integrating part of thecopy control current ICOC. The FET F1, which has a main current path,arranged between the nodes N5 and N6 and a control electrode connectedto the node N5 forms the clamping circuit 82 of FIG. 3. The FET F2,which has a main current path, arranged in parallel with the capacitorC1 and a control electrode connected to the node N6 forms the amplifier83 of FIG. 3.

The limiting circuit which limits the minimum value of the modifiedcurrents IMC of FIG. 6 and FIG. 9 comprises: the current source 80′which supplies the minimum current IMIN from the node N8 to the node N7,the transconductance amplifier TCA7 with transconductance gCOR, thetransconductance amplifier TCA8 with transconductance gHF, thetransconductance amplifier TCA9 with transconductance gLF2, and theFET's F3 and F4. The transconductance amplifier TCA9 receives thevoltage across the capacitor C1 between the non-inverting input and theinverting input, and supplies its output currents to the nodes N7 andN8. Thus, the transconductance amplifier TCA8 supplies the proportionalpart of the copy control current ICOC of FIG. 9 to the nodes N7 and N8,and the transconductance amplifier TCA9 supplies the integrating part ofthe copy control current ICOC. The transconductance amplifier TCA7 has anon-inverting input which receives the output voltage Vo and aninverting input which is connected to a reference voltage which isground, and supplies the correction currents ICRC to the nodes N7 andN8. The FET F3, which has a main current path, arranged between thenodes N7 and N8 and a control electrode connected to the node N7 formsthe clamping circuit 82 of FIG. 9. The FET F4, which has a main currentpath, arranged in parallel with the capacitor C1 and a control electrodeconnected to the node N8 forms the amplifier 83 of FIG. 9.

FIG. 11 shows a block diagram of another embodiment of the current-modecontrolled DC/DC converter in accordance with the invention. FIG. 11shows an adaptation of the prior art converter shown in FIG. 1. Now thecorrection circuit 7 is inserted between the controller 11 and thecomparator 3, while the slope compensation circuit 2 has been left out.The correction circuit 7 receives the control signal ICO and supplies amodified control signal MCO to the comparator 3.

The limiting circuit 8 is added to limit the maximum or minimum value ofthe control signal ICO. Because the control signal ICO is nowrepresentative for the average output current IOA, the limiting circuit8 limits the average output current IOA of the converter.

FIG. 12 shows a block diagram of yet another embodiment of thecurrent-mode controlled DC/DC converter in accordance with theinvention. FIG. 12 shows an adaptation of the prior art converter shownin FIG. 1. Now the correction circuit 7 is inserted between the .current-mode controller 11 and the slope compensation circuit 2. Again,the limiting circuit 8 is added to limit the maximum or minimum value ofthe control signal ICO. The correction circuit 7 receives the controlsignal ICO and supplies a modified control signal MCO to the slopecompensation circuit 2. The slope compensation circuit 2 supplies themodified control signal SCO′ to the comparator 3.

The limiting circuit 8 is added to limit the maximum or minimum value ofthe control signal ICO. Because the control signal ICO is nowrepresentative for the average output current IOA, the limiting circuit8 limits the average output current IOA of the converter.

It should be noted that the above-mentioned embodiments illustraterather than limit the invention, and that those skilled in the art willbe able to design many alternative embodiments without departing fromthe scope of the appended claims.

For example, the current directions all may be inversed. The skilledperson easily understands how to adapt the embodiments shown if PMOSTFET's are replaced by NMOST FET's and the other way around.

In the claims, any reference signs placed between parentheses shall notbe construed as limiting the claim. Use of the verb “comprise” and itsconjugations does not exclude the presence of elements or steps otherthan those stated in a claim. The article “a” or “an” preceding anelement does not exclude the presence of a plurality of such elements.The invention may be implemented by means of hardware comprising severaldistinct elements, and by means of a suitably programmed computer. Inthe device claim enumerating several means, several of these means maybe embodied by one and the same item of hardware. The mere fact thatcertain measures are recited in mutually different dependent claims doesnot indicate that a combination of these measures cannot be used toadvantage.

1. A controller comprising: a comparator for comparing an input signalwith a reference signal to obtain an error signal an integrator forapplying an integrating action on the error signal to obtain a controlsignal the integrator allowing influencing of the integrating action,copy means for supplying a copy control signal being proportional to thecontrol signal determination means for determining whether the copycontrol signal reaches a limit value and influencing means forinfluencing the integrating action to limit the control signal when thecopy control signal reaches the limit value
 2. A controller as claimedin claim 1, wherein the limit value a maximum level and wherein theinfluencing means is arranged for decreasing the integrating action whenthe copy control signal reaches the maximum level
 3. Anintegrating-controller as claimed in claim 1, wherein the limit valueindicates a minimum level and wherein the influencing means is arrangedfor increasing the integrating action when the copy control signalreaches the minimum level
 4. A controller as claimed in claim 1,wherein: the copy means comprises a first current source for supplyingthe copy control signal as a first current to a node the determinationmeans comprises (i) a second current source for supplying apredetermined fixed second current to the node, and (ii) a clampingcircuit for limiting a voltage at the node wherein the first current andthe second current have an opposite polarity, and the influencing meanscomprises an amplifier having an input connected to the node and anoutput connected to an input of the integrator for influencing theintegrating action of the integrator.
 5. A controller as claimed inclaim 4, wherein the integrator comprises an integrating capacitor, andthe output of the amplifier is coupled to the integrating capacitor
 6. Acontroller as claimed in claim 4, wherein the integrator comprises athird current source for supplying the control signal as a third currentbeing determined by a voltage across the integrating capacitor
 7. Acontroller as claimed in claim 6, wherein the amplifier comprises atransistor having a control input coupled to the node and a main currentpath being coupled to the integrating capacitor
 8. A controller asclaimed in claim 4, wherein the clamping circuit comprises a transistorhaving a control input and a main current path coupled to the node, anda voltage source coupled to the control input.
 9. A controller asclaimed in claim 1, wherein the comparator comprises a firsttransconductance amplifier having inputs for receiving the input signaland the reference signal and first outputs to supply the error signalthe integrator comprises a capacitor arranged between the first outputs,and a second transconductance amplifier having inputs coupled across thecapacitor and second outputs supplying the control signal as an outputcurrent the copy means comprises a third transconductance amplifierhaving inputs coupled across the capacitor, and third outputs forsupplying the copy control signal as a copy control current, thedetermination means comprises a current source arranged between thethird outputs and a first transistor arranged between the third outputsto act as a clamp, the current source being arranged for supplying thelimit value as a limit current, and the influencing means comprises atransistor having a main current path arranged between the firstoutputs, and a control input coupled to one of the third outputs forinfluencing a voltage across the capacitor if the copy control currentreaches the limit currently.
 10. A current-mode controlled DC/DCconverter for receiving a power supply input voltage to supply a powersupply output voltage to a load the current-mode controlled DC/DCconverter comprises: the controller as claimed in claim 1 wherein theinput signal is the power supply output voltage an inductor and acontrollable switch being coupled to the inductor for obtaining aperiodically varying inductor current through the inductor a drivecircuit for comparing a sensed signal being representative for theinductor current with the control signal to switch off the controllableswitch when a level of the sensed signal reaches a level of the controlsignal
 11. A current-mode controlled DC/DC converter as claimed in claim10, further comprising a correction circuit for adding to the controlsignal , a correction signal being representative for a differencebetween an original level of the control signal and an average value ofthe inductor current to obtain a modified control signal and wherein thedrive circuit is arranged for comparing sensed signal with the modifiedcontrol signal to switch off the controllable switch when a level of thesensed signal reaches a level of the modified control signal
 12. Acurrent-mode controlled DC/DC converter as claimed in claim 11, furthercomprising means for supplying a copy correction signal beingproportional to the correction signal and wherein: the determinationmeans are arranged for determining whether a sum of the copy controlsignal and the copy correction signal reaches the limit value and theinfluencing means are arranged for limiting the control signal when thesum of the copy control signal and the copy correction signal reachesthe limit value
 13. A current-mode controlled DC/DC converter as claimedin claim 10, wherein: the means for supplying the copy control signalcomprises a first current source for supplying the copy control signalas a first current to a first node the determination means comprises asecond current source for supplying a predetermined fixed second currentto the first node and a clamping circuit for limiting the voltage at thefirst node and the influencing means comprises an amplifier having aninput connected to the first node and an output connected to an input ofthe integrator for influencing the integrating action of the integrator14. A current-mode controlled DC/DC converter as claimed in claim 13,further comprising a second current source for supplying a secondcurrent proportional to the correction current to the first node andwherein the amplifier increases the integrating action when the firstcurrent drops below the sum of the minimum current level and the thirdcurrent
 15. A current-mode controlled DC/DC converter as claimed inclaim 10, wherein: the controller comprises a first current sourcesupplying to a first node a control current being determined by thecontrol signal and wherein the current-mode controlled DC/DC converterfurther comprises a sense circuit for sensing the inductor current tosupply the sensed signal being a sensed current to the first node apolarity of the control current being opposite to a polarity of thesensed current and wherein the drive circuit is coupled to the firstnode for determining when a level of the sensed current reaches a levelof the sum of the control current and the correction current
 16. Anapparatus comprising a current-mode controlled DC/DC converter asclaimed in claim 10, and signal processing circuits for receiving thepower supply output voltage generated by the current-mode controlledDC/DC converter.
 17. An apparatus as claimed in claim 16 being a mobileapparatus comprising a battery for supplying a battery voltage being thepower supply input voltage the current-mode controlled DC/DC converterbeing arranged for converting the battery voltage into the power supplyoutput voltage
 18. A method of controlling comprising: comparing aninput signal with a reference signal to obtain an error signal, applyingan integrating action on the error signal to obtain a control signally,the integrator allowing influencing the integrating action, supplying acopy control signal being proportional to the control signal determiningwhether the copy control signal reaches a limit value and influencingthe integrating action to limit the control signal when the copy controlsignal reaches the limit value